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 NOAA Technical Report ERL 392-WPL 51 
 
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 fc Frequency Spectrum Analyzer 
 
 ff| / for Doppler Lidar 
 
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 M. J. Post 
 
 R. E. Cupp 
 
 R. L. Schwiesow 
 
 October 1976 
 
 U.S. DEPARTMENT OF COMMERCE 
 
 National Oceanic and Atmospheric Administration 
 Environmental Research Laboratories 
 
NOAA Technical Report ERL 392-WPL 51 
 
 t&XjjOSfite 
 
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 Frequency Spectrum Analyzer 
 for Doppler Lidar 
 
 1 
 
 l 
 
 M. J. Post 
 
 R. E. Cupp 
 
 R. L. Schwiesow 
 
 Wave Propagation Laboratory 
 Boulder, Colorado 
 
 October 1976 
 
 U.S. DEPARTMENT OF COMMERCE 
 
 Juanita M. Kreps, Secretary 
 
 National Oceanic and Atmospheric Administration 
 Richard A. Frank, Administrator 
 
 Environmental Research Laboratories 
 Wilmot Hess, Director 
 
Digitized by the Internet Archive 
 
 in 2013 
 
 http://archive.org/details/frequencyspectruOOpost 
 
CONTENTS 
 
 Page 
 
 Abstract 1 
 
 1 . INTRODUCTION 1 
 
 2. SYSTEM CONFIGURATION 2 
 
 3. COMPARISON OF DATA 4 
 
 4. ACKNOWLEDGMENTS 5 
 
FREQUENCY SPECTRUM ANALYZER FOR DOPPLER LIDAR 
 
 M. J. Post, R. E. Cupp, R. L. Schwiesow 
 
 This report describes an electronic apparatus that 
 analyzes Doppler returns from an infrared lidar system. 
 By processing each spectral frequency channel with a 
 100 percent duty cycle rather than with a swept filter 
 analyzer, considerably better S/N is obtained. 
 
 1 . INTRODUCTION 
 
 In Doppler systems, the frequency of the received signal is 
 shifted from the transmitted frequency by an amount proportional 
 to the line-of-sight velocity component of a moving target. 
 Typically, the output signal is the frequency difference between 
 the transmitted and received beams. For our CW lidar, which 
 transmits at a wavelength of 10.59 pm and detects scattering from 
 natural aerosols carried with the wind, this Doppler frequency 
 lies in the range to 20 MHz. 
 
 The received signal for such systems is often weak and inter- 
 mittent, complicating the task of analyzing the signal for its 
 frequency spectrum. Conventional spectrum analyzers scan the 
 signal with a swept narrowband filter, so that a particular fre- 
 quency band (or channel) interrogates the signal only a small 
 percentage of the time, typically 0.5 percent. Unfortunately, in 
 the case of intermittent signals, a given frequency component will 
 be present only sporadically, so that whether or not it is ob- 
 served depends on its timing with respect to the scanning filter 
 system. 
 
 More specifically, conventional analysis techniques (swept 
 filter) lose 11.5 dB of signal-to-noise ratio (S/N) for intermit- 
 tent signals, the square root of the duty cycle. Most of our 
 signals are of this type, since they arise from the infrequent 
 transit of large, natural aerosol "tracers" through the small 
 focal volume of the lidar. The limiting noise for this system is 
 constant "shot noise" generated at the detector. 
 
We developed our analyzer to overcome these signal losses due 
 to low duty cycle. By processing all channels in parallel instead 
 of in succession, it results in 100 percent duty cycle for each 
 channel. The individual channels are then sampled in succession 
 to display the spectra in the conventional manner of intensity 
 versus frequency. Although the design for this analyzer is 
 straightforward and the components are standard, the combination 
 and operation of the elements are novel. No commercial analyzer 
 that we have been able to identify is capable of both such wide 
 bandwidth and such high duty cycle, and to our knowledge, no 
 similar device has been previously reported in the literature. 
 The closest similar processor is a surface acoustic wave device 
 (dispersive delay line) with lower duty cycle, and it is available 
 only on a custom basis. This spectrum analyzer does not have the 
 variable frequency range and bandwidth of the swept-filter type, 
 but its capabilities are not essential to our application. 
 
 2. SYSTEM CONFIGURATION 
 
 The system configuration of our analyzer is depicted in 
 Figure 1. The parallel-processed channels are shown on the far 
 right side. It was decided arbitrarily that 64 discrete channels 
 would yield sufficient frequency resolution for analysis of the 
 Doppler signal. 
 
 Optical 
 Doppler 
 Signal 
 
 Optica 
 Local Oscillator 
 
 Output to 
 Display 
 
 Commi 
 
 Display 
 Trigger 
 
 TIqujki 1. Block diagram o{) VopploA tidcut &pe.c&uim cinalyzzA. 
 
The Doppler signal is shifted upward approximately 51 MHz 
 from the original to 20 MHz signal to reduce both the fractional 
 bandwidth of each channel and the physical size of the processing 
 elements. To accomplish this shift, the a.c. signal from the 
 detector is first amplified and then mixed with a 51-MHz crystal 
 controlled oscillator to produce sum and difference frequencies. 
 The difference frequencies below 51 MHz are rejected by a high- 
 pass filter, while the sum frequencies are amplified and distri- 
 buted in parallel to the bank of processing elements. Distribu- 
 tion is effected with ordinary 75-ohm TV antenna splitters. To 
 prevent overloading the detectors, automatic gain control is 
 applied to the RF signal before distribution under strong signal 
 conditions . 
 
 Each channel consists of a high-frequency detector, a d.c. 
 amplifier, and a near end-fed, shorted, tuned coaxial line, which 
 acts as the filter. This filter resonates at a particular fre- 
 quency and, when that frequency is present in the distributed 
 signal, a resonant level is detected. The filter bandpass is 
 approximately Gaussian, with adjacent filters set to cross over at 
 the 3 dB points. A frequency spike will therefore contribute 
 signals to several adjacent channels and is displayed with consid- 
 erable bandwidth. However, this poses no problem since most 
 velocity spectra in the atmosphere are broad because of flow 
 turbulence . 
 
 Associated with each detector and amplifier is a 0.1-sec 
 integration time constant. Each channel averages the detected 
 signal levels over this period of time. The d.c. amplifiers used 
 in the detection circuits have low thermal drift and a d.c. offset 
 capability to standardize their no-signal levels. Feed and signal 
 pickoff locations, with respect to the shorted end of the line, 
 are critical to maintain desired bandwidth (resonant Q) . Table I 
 shows the empirically determined feed, pickoff, and overall 
 lengths for the first 16 coax-filter elements used, corresponding 
 to the first 16 m/s of the velocity spectrum. We use an applied 
 signal for final trimming to length to insure accuracy. One-inch 
 diameter foam-filled coaxial cable is used to obtain the required 
 
 The 64 amplified detector levels are placed on the terminals 
 of a 64-position electronic commutator. The pole of the commuta- 
 tor is the output of the spectrum analyzer. As the commutator 
 cycles through the 64 channels, an oscilloscope is swept synchron- 
 ously to display the entire spectrum of detected levels as a 64- 
 bar histogram. The commutator may be driven at a variable rate, 
 but it normally cycles through the entire filter bank five times 
 per integration time constant. This rate yields a flicker-free 
 display on the oscilloscope, and permits recording without resolu- 
 tion degradation on a 5-kHz magnetic tape. 
 
TABLE I. 
 
 EMPIRICALLY 
 
 DERIVED PARAMETERS 
 
 OF THE FIRST 
 
 16 
 
 
 RESONANT 
 
 FILTER ELEMENTS 
 
 
 
 Filter No. 
 
 Frequency, 
 
 MHz 
 
 Length (m) 
 
 Feed Tap 
 (cm) 
 
 Pickoff Tap 
 (cm) 
 
 1 
 
 50.340 
 
 
 1.169 
 
 1.98 
 
 7.92 
 
 2 
 
 50.529 
 
 
 1.167 
 
 1.97 
 
 7.90 
 
 3 
 
 50.718 
 
 
 1.160 
 
 1.96 
 
 7.87 
 
 4 
 
 50.907 
 
 
 1.159 
 
 1.96 
 
 7.85 
 
 5 
 
 51.096 
 
 
 1.154 
 
 1.94 
 
 7.82 
 
 6 
 
 51.285 
 
 
 1.146 
 
 1.95 
 
 7.80 
 
 7 
 
 51.473 
 
 
 1.144 
 
 1.94 
 
 7.75 
 
 8 
 
 51.662 
 
 
 1.139 
 
 1.93 
 
 7.72 
 
 9 
 
 51.851 
 
 
 1.136 
 
 1.93 
 
 7.70 
 
 10 
 
 52.040 
 
 
 1.131 
 
 1.92 
 
 7.67 
 
 11 
 
 52.229 
 
 
 1.127 
 
 1.91 
 
 7.65 
 
 12 
 
 52.418 
 
 
 1.124 
 
 1.91 
 
 7.62 
 
 13 
 
 52.606 
 
 
 1.123 
 
 1.90 
 
 7.59 
 
 14 
 
 52.795 
 
 
 1.116 
 
 1.89 
 
 7.57 
 
 15 
 
 52.984 
 
 
 1.111 
 
 1.88 
 
 7.54 
 
 16 
 
 53.173 
 
 
 1.106 
 
 1.88 
 
 7.52 
 
 3. COMPARISON OF DATA 
 
 The superiority of this analyzer over the best similar com- 
 mercial unit is demonstrated in Figure 2. These oscilloscope 
 traces (linear intensity vs. frequency) were taken simultaneously 
 for the conventional spectrum analyzer (a and b) and for our 
 parallel processing analyzer (c and d) . Common to both analyzers 
 was a 7-dB noise figure preamplifier. These spectra were obtained 
 for weak signal conditions (backscattering from "clear air" aero- 
 sols at 100 m range). The conventional analyzer was operated at 
 100 ms per scan to be consistent with the 100-ms time constant of 
 our parallel filter system. At least 7 dB of the 11.5 dB loss in 
 S/N ratio inherent in the conventional analysis was recovered in 
 our parallel analysis. 
 
 Trace 2(b) shows unity S/N, while trace 2(d) has approxi- 
 mately 8 dB S/N. The apparent noise on the higher frequency 
 portion of the spectrum in 2(c) and 2(d) is long-term differential 
 thermal drift, which may be subtracted from the data mathemati- 
 cally or eliminated with improved electronic design. The multiple 
 traces reflect the intermittency of the scattered signal, smoothed 
 by the 0.1-sec time constant of the detector. 
 
Swept Filter 
 Analyzer 
 
 (c) 
 
 Parallel Processing 
 Analyzer 
 
 Output -3 
 Volts 
 
 i — i — i — i — i — i — i — i — r 
 
 .2 .4 .6 .8 1.0 1.2 1.4 1.6 1.8 
 MHz 
 
 n — i — i — i — i — i — i — i — T 
 
 .2 .4 .6 .8 1.0 1.2 1.4 1.6 1.8 
 MHz 
 
 FxdguAe 1. SpudtAd psioceAAdd by bojvlaJL [a,b] and paxaltit [c,d] 
 
 <matijz&u> . 
 
 The frequency stability of the filter bank has been tested 
 over a period of several months and has been excellent. Still to 
 be performed is extensive field calibration of the entire analyzer 
 to determine the effects of temperature, humidity, or other oper- 
 ating conditions on the produced spectra. However, with only re- 
 zeroing of the individual d.c. amplifiers for changes in ambient 
 temperature, the analyzer has performed satisfactorily in the 
 field. Therefore, if versatility is not required, this type of 
 analyzer offers specific advantages over conventional analyzers, 
 in terms of both signal-processing efficiency and cost. 
 
 4 . ACKNOWLEDGMENTS 
 
 The authors wish to acknowledge the help provided by G. M. 
 Lerfald and H. L. Ericson in the design and construction of the 
 frequency spectrum analyzer. 
 
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